Sixties-Style 40W Audio Amplifier

In the early 1960s RCA brought out a transistor that was to become truly legendary: the 2N3055. With a pair of these devices, you could put together an audio power amplifier that could deliver a healthy 40 W into 8 Ω. The circuit described here is fully in tune with the spirit of that era. For example, there are only seven active components in each channel, which reflects the design simplicity typical of that era (and actually a timeless quality). This ‘retro’ power amplifier pumps 45 W into 8 Ω with an input signal level of 0.5 Vrms. It works as follows: the input signal is applied to the base of T1, while negative feedback from the output, attenuated by voltage divider R5/R6, is applied to the emitter of T1. The collector current of T1, which is proportional to the difference between the input and feedback signals, is fed to the base of T2.

Amplifier Circuit Diagram:

This transistor draws its operating current through R8 and R9 and provides voltage gain. Capacitor C6 is a bootstrap capacitor that hold the voltage across R9 nearly constant, so that the current through R9 is independent of the amplifier output signal level in the audio band. Transistors T4–T7 form a quasi-complementary push-pull output stage. In the early 1960s, there simply wasn’t any PNP transistor available that was truly complementary to the 2N3055. Designers came up with an ingenious way to get round this problem, which was to use a complementary Darlington pair consisting of a PNP driver transistor and an NPN power transistor. The schematic diagram clearly illustrates what is meant by a quasi-complementary push-pull output stage. Diode D1 provides balanced biasing for the output stage, which helps reduce distortion.

The operating point of the output stage is set and stabilized by transistor T3, which for this reason should be thermally coupled to the output transistors. The amplifier is powered from a single supply voltage at approximately 65 V, which is also ‘typical sixties’. Capacitor C1, with a value of 4700 μF, transfers the signal from the output stage to the load and provides a bit of protection for the speaker in case one of the transistors fails. The amplifier does not have output current limiting. Although this is not a critical shortcoming, a certain amount of caution is advisable. The only protection in this regard is provided by a slow-acting 1.6-A fuse in the supply line, which is intended to limit the damage if anything goes wrong.

Power Supply Circuit Diagram:

The power supply consists of a transformer, a bridge rectifier, four small capacitors and a 4700 μF electrolytic capacitor. This is enough to power a two-channel stereo amplifier. The LED is a power-on indicator and is intended to be fitted on the front panel. Assembling the circuit is very straightforward. Transistors T3, T4 and T5 should be fitted with heat sinks suitable for a TO126 package and with a thermal resistance less than 20 K/W. Transistors T2, T6 and T7 should all be fitted on a single heat sink with a thermal resistance of 2 K/W or less, using insulating washers and thermal paste. Before applying power to the circuit for the first time, set P2 to its maximum value, temporarily replace the fuse with a 47 Ω, 5 W resistor, and connect a voltmeter across R17.

Then switch on the power. The voltmeter should indicate 0 V. Now carefully adjust P2 until the voltmeter reads 15 mV, which corresponds to a quiescent current of 50mA. Then switch off the power and install the fuse in place of the power resistor. After this, check the voltage across R17 again (with the power on) and if necessary adjust it to 15mV. This is fun DIY project, cheap and unpretentious. Nevertheless, the sound quality of this amplifier is respectable. The distortion level gives no grounds for complaint. Of course, it’s not a figure with an incredible number of zeros after the decimal point, but the idea here is to brush up on sixties technology.



Author: Joseph Kreutz (Germany) - Copyright: Elektor Electronics 2011

 

 

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Voltage Limiter For Guitar Amplifiers

Guitar amplifiers using output devices such as the TDA7293 (100 W) or LM3886 (68 W) are surprisingly often damaged as a result of excessive supply voltage in the quiescent state. The transformers are often used so close to their specification that this problem can even be caused by a high mains input voltage. In most countries the domestic AC outlet voltage is permitted to rise as high 10 % above the nominal (published) value. Since replacing the transformer is not an attractive proposition, the author developed a relatively simple electronic solution to the overvoltage problem: a voltage limiter for the symmetric supply to the amplifier. The circuit is based on the classical voltage regulator arrangement of a Zener diode connected to the base of a pass transistor.

However, in this version we replace the conventional bipolar transistor with a power MOSFET. The circuit is symmetrical with respect to the negative and positive supplies, and so we shall only describe the positive half. The input voltage (at most 50 V) supplies the chain of Zener diodes D1, D2 and D3 via resistor R3. The resistor limits the current through the Zener diodes to about 5mA. The series connection of Zener diodes has the advantage that their dissipation is divided among them, as well as giving more options for the total voltage drop by judicious selection of individual components. The sum of the diode voltages (39 V with the values given) must be greater than the desired limiting output voltage by the gate-source voltage of the MOSFET.

Circuit diagram:

C1 smooths the voltage across the Zener diode chain. The circuit therefore not only limits the voltage, but also reduces the ripple (hum component) on the supply. The gate of the HEXFET is driven via R1. In conjunction with C4, this prevents the FET from oscillating. Without any load the output voltage is rather higher than expected. With a small load, such as that presented by the output stage in its quiescent state, it falls to the desired value. The circuit then does not provide regulation of the output voltage, but rather a stabilization function. The operation of the negative half of the circuit is identical to that of the positive half apart from the polarity of the voltages, and hence a P-channel MOSFET must be used there.

It is worth noting that there is a relatively large degree of variation (up to a few volts) in the gate-source voltage of the HEXFETs used. This can be compensated for by selecting the Zener diodes in the chain and the current through them, but for most applications the exact voltage at which limiting begins to occur will not be critical. The HEXFETs must be provided with adequate cooling. If possible, they can be attached to the heatsink already present in the amplifier; otherwise, a separate heatsink will be required. A thermal rating of 2.5 K/W will be suitable.



Author: Alfred Rosenkränzer (Germany) - Copyright: Elektor Electronics 2011

 

 

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Random Number Generator Based Game

This electronic game is simulation of one-arm bandit game. Electronics hobbyists will find it very interesting. When toggle switch S1 is in ‘run’ position, all segments of 7-segment displays (DIS1 through DIS3) will light up. On turning toggle switch S1 from ‘run’ to ‘stop’ position, displayed digits will continue advancing and the final display is unpredictable. Thus the final number displayed in DIS1 through DIS3 is of random nature. The speed with which the number in 7-segment display keeps changing on flipping switch S1 from ‘run’ to ‘stop’ condition slowly decays before stopping with a random number display. To play this game, one has to obtain three identical numbers in displays DIS1 through DIS3.

The contestant would score 1 (one) point if he manages to get a final display of ‘000’, 2 points for getting ‘111’ display, 3 points for ‘222’,… and so on—up to ten points for ‘999’. He should try to score maximum possible points in fixed numbers of attempts (say, 20 to 25 attempts). Apart from using this circuit as a game for entertainment, one can use it as random number generator for any other application as well. The decay time with the given component values is around 15 seconds before the display could stop at a final random number.

Circuit diagram:
The circuit comprises clock oscillator built around NE555 timer IC4, three-stage clock pulse counter built using three CD4033 ICs (IC1 to IC3), and three 7-segment LED displays (DIS1 to DIS3). In clock oscillator circuit, NE555 timer IC4 is used in a similar way as a free-running astable multivibrator, the only difference being the additional capacitor C1 introduced between pin No. 7 of IC4 and junction of resistors R22 and R24. When toggle switch S1 is in ‘run’ position, both terminals of capacitor C1 are shorted by switch S1 and timer IC4 works as a free-running astable multivibrator. The operating frequency is in the vicinity of 35 kHz, determined by the value of timing components.

When toggle switch S1 is flipped from ‘run’ to ‘stop’ position, capacitor C1 is introduced in the discharge path of pin No. 7 of IC4 and junction of resistors R22 and R24. At the same time, capacitor C4 comes in parallel with timing capacitor C3 to change the operating frequency of the astable from around 35 kHz to around 65 Hz. Now capacitor C1 slowly starts charging as it is connected in the discharge path of the timing capacitors C3 and C4. The clock frequency of IC4 gradually reduces and after 15 seconds, when capacitor C1 is sufficiently charged, the oscillating frequency gradually drops and finally it stops oscillating. Thus, pin 3 of IC4 becomes low.

Second part of the circuit comprises three cascaded ICs, IC1 through IC3 (CD4033 decade upcounter cum 7-segment decoder). In conjunction with three 7-segment displays (DIS1 to DIS3), these form a 3-digit clock counter. The clock counting speed is dependant upon the clock pulse frequency of IC4. It is connected to clock input pin 1 of IC1 while chip enable pin 2 of IC1 to IC3 are held low. Thus all clock counter ICs advance by 1 for every positive clock transition. Reset pin 15 of all counter ICs is held low through resistor R25. Thus reset facility is not used in this circuit. Due to persistence of vision, one can not distinguish 0-9 counting in DIS1 to DIS3 when the clock frequency is high.

All 7-segment displays appear to show digit 8, while the red LED1 remains lit continuously, indicating clock counter is in running condition. On sliding toggle switch S1 from ‘run’ to ‘stop’ position, the counting speed of individual digits falls immediately due to the clock frequency changing to around 65 Hz. Now, the counting speed will be 65 Hz for DIS3, 6.5 Hz for DIS2, and 0.6 Hz for DIS1. This speed of individual digit counting slowly decays, until the counter stops and LED1 stops blinking, and the final count (random numbers) are displayed in DIS1, DIS2, and DIS3.

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High Low Voltage Cutout Without Timer

This inexpensive circuit can be connected to an air-conditioner/fridge or to any other sophisticated electrical appliance for its protection. Generally, costly voltage stabilizers are used with such appliances for maintaining constant AC voltage. However, due to fluctuations in AC mains supply, a regular ‘click’ sound in the relays is heard. The frequent energisation/de-energisation of the relays leads to electrical noise and shortening of the life of electrical appliances and the relay/stabilizer itself. The costly yet fault-prone stabiliser may be replaced by this inexpensive high-low cutout circuit with timer.

The circuit is so designed that relay RL1 gets energised when the mains voltage is above 270V. This causes resistor R8 to be inserted in series with the load and thereby dropping most of the voltage across it and limiting the current through the appliance to a very low value. If the input AC mains is less than 180 volts or so, the low-voltage cut-off circuit interrupts the supply to the electrical appliance due to energisation of relay RL2. After a preset time delay of one minute (adjustable), it automatically tries again. If the input AC mains supply is still low, the power to the appliance is again interrupted for another one minute, and so on, until the mains supply comes within limits (>180V AC).

Circuit diagram:
The AC mains supply is resumed to appliance only when it is above the lower limit. When the input AC mains increases beyond 270 volts, preset VR1 is adjusted such that transistor T1 conducts and relay RL1 energises and resistance R8 gets connected in series with the electrical appliance. This 10-kilo-ohm, 20W resistor produces a voltage drop of approximately 200V, with the fridge as load. The value and wattage of resistor R8 may be suitably chosen according to the electrical appliance to be used. It is practically observed that after continuous use, the value of resistor R8 changes with time, due to heating. So adjustment of preset VR1 is needed two to three times in the beginning.

But once it attains a constant value, no further adjustment is required. This is the only adjustment required in the beginning, which is done using a variac. Further, the base voltage of transistor T2 is adjusted with the help of preset VR2 so that it conducts up to the lower limit of the input supply and cuts off when the input supply is less than this limit (say, 180V). As a result, transistor T3 remains cut off (with its collector remaining high) until the mains supply falls below the lower limit, causing its collector voltage to fall. The collector of transistor T3 is connected to the trigger point (pin 2) of IC1. When the input is more than the lower limit, pin 2 of IC1 is nearly at +Vcc.

In this condition the output of IC1 is low, relay RL2 is de-energised and power is supplied to the appliance through the N/C terminals of relay RL2. If the mains supply is less than the lower limit, pin 2 of IC1 becomes momentarily low (nearly ground potential) and thus the output of IC1 changes state from ‘low’ to ‘high’, resulting in energisation of relay RL2. As a result, power to the load/appliance is cut off. Now, capacitor C2 starts charging through resistor R6 and preset VR3. When the capacitor charges to (2/3)Vcc, IC1 changes state from ‘high’ to ‘low’. The value of preset VR3 may be so adjusted that it takes about one minute (or as desired) to charge capacitor C1 to (2/3)Vcc.

Relay is now de-energised and the power is supplied to the appliance if the mains supply voltage has risen above the lower cut-off limit, otherwise the next cycle repeats automatically. One additional advantage of this circuit is that both relays are de-energised when the input AC mains voltage lies within the specified limit and the normal supply is extended to the appliance via the N/C contacts of both relays.

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Touch-Select Audio Source

Often you need to connect output from more than one source (preamplifier) such as tape recorder/player and CD (compact disc) player to audio power amplifier. This needs disconnecting/connecting wires when you want to change the source, which is quite cumbersome and irritating. Here is a circuit that helps you choose between two stereo sources by simple touch of your hand. This circuit is so compact that it can be fixed within the audio power amplifier cabinet and can use the same power supply source. The circuit uses just two CMOS ICs and a few other componenets. The ICs used are MC14551/CD4551 (quad 2-channel analogue multiplexer) and CD4011 (quad 2-input NAND gate).

When touch-plate S1 is touched (its two plates are to be bridged using a fingertip), gate N1 output (IC1, pin 3) goes high while the output of gate N2 at pin 4 goes low. This causes selection of CD outputs being connected to the power amplifier input, which is indicated by lighting of LED1. When touch-plate S2 is touched, the outputs of gates N1 and N2 toggle. That is, IC2 pin 3 is pulled ‘low’ while its pin 4 goes ‘high’. This results in selection of tape recorder outputs being connected to the input of power amplifier. This is indicated by lighting of LED2. Pin 9 is the control pin of IC2.

In the circuit, the state of multiplexer switches is shown with pin 9 ‘high’ (CD source selected). When pin 9 is pulled ‘low’, all the switches within the multiplexer change over to the alternate position to select tape player as source. Note. Although one can connect pin 7 (VEE) of IC2 to ground, but for operation with preamplifier signals going above and below ground level, one must connect it to a negative voltage (say, –1V to –1.5V) to avoid distortion.

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Electric Guitar Preamp, Mixer and Line Driver

Depending on its design an electric guitar may have anything from one to six pickup elements. Classic (acoustic) guitars could also benefit from one or more retro-fitted pickups. Each pickup has a specific sound depending on the type of sensor and the location on the instrument. When a guitar has more than one pickup these can be connected together with or without additional components. However it is preferable for each pickup signal to be buffered individually. These buffered and possibly amplified signals should be level-adjusted in order to produce the desirable effect (or ‘sound’). After that they are mixed and sent to the next stage of the audio processing equipment.

Circuit diagram:

Electric Guitar Preamp, Mixer and Line Driver Circuit Diagram

Most guitarists agree that pickup elements cannot drive cables longer than about 6 feet without risking significant signal degradation. Guitar pickups typically require a load resistance above 50 kΩ and sometimes higher than 200 kΩ, hence a preamplifier/buffer is often inserted, whose main function is not high gain but to enable cables between 10 and 30 feet to be connected representing a capacitance between 90 and 180 pF/m. In the circuit shown here, each pickup has its own input buffer with a transistor configured as an emitter follower. Each stage has a gain slightly lower than unity. This is not an issue because most pickups provide significant signal levels, typically well over 200 mVpp.

The input resistance of the first stage exceeds 200 kΩ, which is appropriate for most inductive pickups on the market. If higher input resistance is needed the 1-MΩ resistors marked with asterisks could be omitted, and the 720-kΩ ones may be increased to 1.2 – 1.5 MΩ. This will raise the stage’s input resistance to around 500 kΩ. To ensure the highest possible undistorted signal can be developed at the output of the first stages, the collector-emitter voltage (VCE) of T1–T4 should be about half the supply voltage. It is important for the first transistor in the buffer to have low noise and high DC gain.

The types BC549C and BC550C and the venerable BC109C are perfectly suitable in this respect while the BC546C, BC547C and BC548C may also be considered. The buffered signal from each pickup is adjusted with a potentiometer and sent to the summing circuit of the mixer. The next active element is an audio operational amplifier type NE5534 or NE5534A (IC1), which provides the required amount of signal buffering. The 5534(A) has low noise, low distortion and high gain. It can drive a 600 Ω line when necessary, but the preferred load is above 2 kΩ. Its amplification is adjustable between 3 and 10 with feedback potentiometer P5. At higher values of the gain some limiting and distortion of the output signal is ‘achieved’, which may well be a desirable side effect.

The maximum undistorted amplitude of the output signal depends on the supply voltage. If higher gain is needed the value of P5 may be increased to 470 kΩ. Output K7 has a volume control potentiometer (P6), which could be omitted if not used or required. Both outputs K6 and K7 are capable of driving 600 Ω loads including high-impedance headphones. The circuit is simple to test and adjust, as follows:

1. check that VCE on T1–T4 is approximately half the supply voltage;
2. with no input signal, adjust trimpot P7 for about half the supply voltage at the output of IC1. If precise regulation of the opamp’s output offset is not required P7 may be omitted and R17 connected to the junction of R18 and R19.

The supply voltage is between 12 V and 24 V. It is possible to run the unit off a 9 V power supply but the lower supply voltage will limit the output amplitude and gain. The current consumption from a 9 V battery is typically 10 mA. Two 9 V batteries connected in series is the preferred solution. The undistorted output amplitude is up to 6 Vpp at a 12 V supply with 2 kΩ loads at the outputs. The unit’s frequency band exceeds 20 Hz – 20 kHz. Distortion and noise were found to be negligible in view of the application.

Author: Petre Tzvetanov Petrov (Bulgaria) - Copyright: Elektor Electronics 2011

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Audio Level Adapter

The problem that this circuit is designed to solve appeared when the author was installing a new radio in his Audi A3. The new radio had four outputs for loudspeakers and a line-level output for a subwoofer. However, the A3 as delivered from the factory already has an amplifier for the rear loudspeakers, as well as the pre-installed subwoofer, in the boot space. The original Audi radio therefore has only line-level outputs for the rear loudspeakers. So, to replace the original radio without making other changes to the installed amplification system, he needed to convert the outputs of the new radio corresponding to the rear loudspeakers into line level outputs.

Most of the commercially-available adapters to do this job contain small transformers for galvanic isolation. These introduce phase shifts and create a certain amount of distortion, which the author was keen to minimize. The result is this simple adapter circuit that does not employ a transformer. The outputs of most radios available today have a differential (bridge-type) push-pull output stage. There is thus no ground output, just two outputs per channel with a 180 ° phase difference between them. If the outputs are each connected to a common point via a 100 Ω resistor, that point becomes a virtual ground.

Circuit diagram:

Audio Level Adapter Circuit Diagram

The ground is relatively stable as (in the stereo case) it has an impedance of 25 Ω. Each output driver is seeing a 200 Ω load: if the amplifier is rated for 50 W output into a 4 Ω load this means that each resistor will dissipate less than 0.5 W. Hence 1 W rated resistors will be more than adequate, especially in view of the fact that typical music has a crest factor of at least five. Even a small DC offset from the virtual ground is not a problem, as most modern amplifiers feature differential inputs or at least allow the ground connection of an input to float. To reduce the signals to line level, each has to be connected to a potential divider: a multi-turn preset potentiometer is ideal.

The author used a linear 10 kΩ trimmer to reduce the output voltage level from up to about 12 Vpp to around 2 V to 3 V. This latter level is suitable for the input to a power amplifier. An appropriate trimmer setting can be found by ear, adjusting the volume of the rear speakers for the desired balance. There is no need for a printed circuit board for this project. The 1 W resistors can be soldered directly to the connections of the multi-turn presets, and so the whole thing can be assembled ‘in the air’ and shrouded in heat-shrink tubing. The circuit can then be tucked away in the space behind the radio itself.

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Video Switch for Intercom System

Nowadays a lot of intercom units are equipped with video cameras so that you can see as well as hear who is at the door. Unfortunately, the camera lens is perfectly placed to serve as a sort of support point for people during the conversation, with the result that there’s hardly anything left see in the video imagery. One way to solve this problem is to install two cameras on the street side instead only one, preferably some distance apart. If you display the imagery from the two cameras alternately, then at least half of the time you will be able to see what is happening in front of the door.

Circuit diagram:

Video Switch for Intercom System Circuit Diagram

Thanks to the video switch module described here, which should be installed on the street side not too far away from the two cameras, you need only one monitor inside the house and you don’t need to install any additional video cables. Along with a video switch, the circuit includes a video amplifier that has been used with good results in many other electronics projects, which allows the brightness and the contrast to be adjusted separately. This amplifier is included because the distance between the street and the house may be rather large, so it is helpful to be able to compensate for cable attenuation in this manner.

The switch stage is built around the well known 4060 IC, in which switches IC2a and IC2d alternately pass one of the two signals to the output. They are driven by switches IC2b and IC2c, which generate control signals that are 180 degrees out of phase. The switching rate for the video signals is determined by a clock signal from an ‘old standby’ 555 IC, which causes the signals to swap every 2 seconds with the specified component values. Naturally, this circuit can also used in many other situations, such as where two cameras are needed for surveillance but only one video cable is available.

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Generation of 1-Sec Pulses Spaced 5-Sec Apart

This circuit using a dual-timer NE556 can produce 1Hz pulses spaced 5 seconds apart, either manually or automatically. IC NE556 comprises two independent NE555 timers in a single package. It is used to produce two separate pulses of different pulse widths, where one pulse initiates the activation of the second pulse. The first half of the NE556 is wired for 5-second pulse output. When slide switch S2 is in position ‘a’, the first timer is set for manual operation, i.e. by pressing switch S1 momentarily you can generate a single pulse of 5-second duration. When switch S2 is kept in ‘b’ position, i.e. pins 6 and 2 are shorted, timer 1 in NE556 triggers by itself.

The output of the first timer is connected to trigger pin 8 of second timer, which, in turn, is connected to a potential divider comprising resistors R4 and R5. Resistor R1, preset VR1, resistor R2, preset VR2, and capacitors C2 and C5 are the components determining time period. Presets VR1 and VR2 permit trimming of the 5-second and 1-second pulse width of respective sections. When switch S2 is in position ‘a’ and switch S1 is pressed momentarily, the output at pin 5 goes high for about 5 seconds. The trailing (falling) edge of this 5-second pulse is used to trigger the second timer via 0.1µF capacitor C6.

Circuit diagram:This action results in momentarily pulling down of pin 8 towards the ground potential, i.e. ‘low’. (Otherwise pin 8 is at 1/2 Vcc and triggers at/below 1/3 Vcc level.) When the second timer is triggered at the trailing edge of 5-second pulse, it generates a 1-second wide pulse. When switch S2 is on position ‘b’, switch S1 is disconnected, while pin 6 is connected to pin 2. When capacitor C is charged, it is discharged through pin 2 until it reaches 1/3Vcc potential, at which it is retriggered since trigger pin 6 is also connected here. Thus timer 1 is retriggered after every 5-second period (corresponding to 0.2Hz frequency). The second timer is triggered as before to produce a 1-second pulse in synchronism with the trailing edge of 5-second pulse.

This circuit is important wherever a pulse is needed at regular intervals; for instance, in ‘Versatile Digital Frequency Counter Cum Clock’ construction project published in EFY Oct. ’97, one may use this circuit in place of CD4060-based circuit. For the digital clock function, however, pin 8 and 12 are to be shorted after removal of 0.1µF capacitor and 10-kilo-ohm resistors R4 and R5.

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