Power Diode For Solar Power Systems

Apart from the sun, solar power systems cannot work without a reflow protection diode between the solar panel and the energy store. When current flows into the store, there is a potential drop across the diode which must be written off as a loss in energy. In the case of a Schottky diode, this is not less than 0.28 V at nominal current levels, but will rise with higher ones. It is clear that it is advantageous to keep the energy loss as small as possible and this may be achieved with external circuitry as shown in the diagram. The circuit is essentially an electronic switch consisting of a high precision operational amplifier, IC1a, a Type OP295 from Analog Devices, and a MOSFET, T1.

This arrangement has the advantages over a Schottky diode that it has a lower threshold voltage and the lost energy is not dissipated as heat so that only a small heat sink is needed. When the potential at the non-inverting input of the op amp, which is configured as a comparator, rises above that at the inverting input, the output switches to the operating voltage. The transistor then comes on, whereupon light-emitting diode LD1 lights. Diode D3 clamps the inputs of IC1a so that the peak input voltage cannot be greater than half the threshold voltage, provided the values of R3 and R4 are equal.


The op amp provides very high small-signal amplification, a small offset voltage, and consequent fast switching. The MOSFET changes from on to off state and vice versa at drain -source voltages in the microvolt range. In the quiescent state, when UDS is 0 V, the transistor is on, so that LD1 lights. The operating voltage (C–A) may be between 5 V (the minimum supply for the op amp and the input control potential, UGS, of the transistor) and 36 V (twice the zener voltage of D1). Zener diode D1 protects the MOSFET against excessive voltages (greater than ±20 V). Diode D3 and resistors R3 and R4 halve the potential across the inputs of the op amp.

This ensures that operation with reversed or open terminals is harmless. The substrate diode of the MOSFET is of no consequence since it does not become forward biased as long as the forward voltage, USD, of the transistor is held very low. The on -resistance, RSD(on), of the transistor is only 8 mΩ and the transistor can handle currents of up to 75 A. When the nominal current is 10 A, the drop across the on-resistance is 80 mV, resulting in an energy loss of 0.8W. This is low enough for a SUB type with a TO263-SMD case to be used without heat sink. When the current is 50 A, however, it is advisable to use a SUP type with a TO220 case and a heat sink since the transistor is then dissipating 12.5 W.

Even then, the voltage drop, USD = 0.32 V is significantly lower than that across a Schottky diode in the same circumstances. Moreover, owing to the high precision of IC1a, a number of transistors may be used in parallel. The circuit proper draws a current of 150 µA when only one of the op amps in the OP295 is used. An even lower current is drawn by the alternative Type MAX478 from Maxim. However, the differences between these two types are only relevant in the low current and voltage ranges. Both have rail-to-rail outputs that set the control voltage accurately even at very low operating voltages.

This is important since the switch-on resistance of MOSFETs is not constant: t drops significantly with increasing gate potentials and decreasing temperature. A experimental circuit may use an LM358 op amp and a Type BUZ10 transistors, but these components do not give the excellent results just described.

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General-Purpose NiCd Battery Charger

There is a wide variety of NiCd (nickel-cadmium) battery chargers on the market, but there are not many that can work from an in car 12 V cigar lighter. Such a charger would, for instance, be of interest to campers and caravanners who do not have a 230 V a.c. mains supply available. To satisfy the needs of these users, a charger could be designed for operation from the cigar lighter, but it is, of course, of far greater interest if it could also work from the domestic mains supply. Furthermore, it would also be very useful if a number of cells, say, 1 to 4, of different format could be charged simultaneously.

Lastly, another benefit would be if the charger would automatically switch off once the battery or cells have been charged fully. The charger described in this article does all that: it accommodates batteries or cells Type R6 and R14. Switching off after a period of 2 h 30 m, 5 h, or 10 h is arranged by 3-way switch S1. The 2 h 30 m period is for charging Type R6 batteries (1/2 charge), the 5 h period for fully charging Type R6 batteries or half charging Type R14 batteries, and the 10 h period for fully charging Type R14 batteries. Light-emitting diode D1 lights when charging is taking place. Charging after the set period has elapsed can be continued, if so desired, only by switching the supply off and then on again.

The time periods are determined by counters IC1 and IC2, Type 4060 and 4020 respectively. The 4060 has an integral oscillator, whose frequency is set to 932 Hz with preset P1 and the aid of a frequency meter. For various reasons, such as the values of the components used and parasitic elements, the oscillator itself operates at a slightly higher frequency – of the order of 1 kHz. The frequency of the signal at the wiper of P1 is divided by 214, so that the frequency of the signal at Q13 of IC1 is 0.056 Hz, equivalent to a pulse every 17.6 s. The signal at Q13 is applied to the input, pin 10, of IC2. When switch S1 is in position 2 h 5 m (output Q10 of IC2), the divisor should be 210 (1024).

However, contrary to what these figures indicate, the time period stops at half that at output Q10. To obtain a charging period of 2 h 30 m, that is, 9,000 seconds, which should correspond to half a period at output Q9 of IC2, the oscillator period must be 9000×2/16.7×106=1.073 ms, which corresponds to a frequency of 932 Hz as mentioned earlier. On power-on, only counter IC2 is reset, since an error of a few seconds that may arise in IC1 is of no significance. This arrangement simplifies the design. When the time set has elapsed, that is, charging is finished, diode D1 goes out.

The charging current is fixed by darlington transistor T3, which is a classical design of a current source with negative feedback. The transistor tends to hold its emitter potential at 1.3 V, but this requires the aid of a zener diode, D2. In this type of design, the thermal stability is, in fact, totally acceptable, because the temperature of the zener diode, in view of the small current this draws and its consequent low temperature rise, hardly affects the charging current Transistor T1 is either on or off and serves to power the on/off indicator LED. It is needed to prevent an overload on the output of counter IC1 if this would be required to absorb the total current (about 7mA) drawn by the diode.

Transistor T2 discontinues the charging when the time set by S1 has elapsed by earthing the base of darlington T3. Diodes D3–D14 are connected in threesomes across the terminals of the batteries to be charged: D3–D5 across those of battery Bt1, D6–D8 across those of Bt2, and so on. Diode D15 prevents the batteries to be charged from being discharged when the supply fails. When the charger is used in a vehicle, additional precautions should be taken to ensure that any spurious surges on the vehicle power lines do not adversely affect the charger’ s operation. The battery holder should be one that can accommodate four size R6 (AM3; MN1500; SP/HP7; mignon) or R14 AM2; MN1400; SP/HP11; baby) batteries.

The length of these batteries, but not their diameter, is the same (about 45 mm). When the charger is used at home, it may be powered via a suitable 15V mains adaptor. It draws a current of about 150mA. A final word of warning: it is possible for batteries to be connected to the charger with incorrect polarity. This may result in a very large discharge current and even destruction of the battery. It is, therefore, imperative to verify the correct polarity of the battery before inserting it into the holder.

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Absolute-Value Meter With Polarity Detector

This circuit breaks an input voltage signal down into its components: (1) the absolute value and (2) the polarity or ‘sign’ (+ or –). It will handle direct input voltages as well as alternating voltages up to several kHz. With a supply voltage of ±9 V, the input level should remain below ±6V. The circuit consists of two sections, each having its own function. Operational amplifiers IC1a and IC1b form a full-wave rectifier, its output terminal supplying the absolute value of the input signal, while operational amplifiers IC1c and IC1d examine the polarity of the input voltage. For negative input voltages, the output of IC1a goes high.

Consequently D2 is reverse-biased so that IC1a has no effect on the rest of the circuit. IC1b then acts as an inverter because its amplification is (–R5/R3) or –1. Thus, the output voltage is positive. For positive input voltages, D2 conducts and the amplification of IC1a is -1. The output voltage is then determined by the sum of currents that flow through R3 and R4. Taking into account the polarities and the value of all resistors, the overall amplification becomes

–R5/R3 + (–R5/R4) ↔ (–R2/R1) = –1 + 2 = 1

This means that the value of the output voltage at the output terminal is the same as the input voltage, but the polarity is always positive. The accuracy of the rectification process is determined by the accuracy of resistors R1-R4; close-tolerance (1%) types are recommended. At low input voltages (smaller than 20 mV), the input offset voltage of the operational amplifiers may introduce significant errors. If this is the case, use individual operational amplifiers instead of an array of four (TL061, TLC271, AD548, ...), because they have pins for offset voltage compensation. Alternatively, use an operational amplifier with a low offset voltage like the OP07. In the polarity detector IC1c acts as a comparator, with a certain amount of positive feedback due to R7 and R8.

This feedback causes a hysteresis of 20mV that prevents oscillation when the input voltage changes slowly. IC1d is an ordinary inverter. For input voltages above 10mV, the SIGN output terminal will swing to almost the positive supply. When the input voltage is below –10mV, the SIGN terminal drops low, almost to the negative supply voltage. For input voltages between these two thresholds, the output voltage is well defined, too, because it stays at its previous level. This circuit is the perfect complement to the ‘+/– voltage on bargraph display’ circuit discussed elsewhere in this blog.

The |Uin| and SIGN outputs of the present circuit may be directly connected to Uin and CONTROL IN inputs of the bi-directional bar display. The ±6 V sign indicator signal may be used as the control voltage for the +/– voltage display as long as the reference voltage remains smaller than 3 V. Although presented as a pair, both circuits may of course be used individually for other purposes.

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Multiple Continuity Tester

The continuity tester is a handy adjunct to an ohmmeter. The unit or component whose continuity is to be checked is connected between terminals E1 and E2 (which may be probes or croc clips). The test current then flowing through the unit/component on test causes a potential drop across resistor R2, which is applied to the non-inverting input of buffer IC2. The output of the op amp is applied to transistor T1, in the emitter circuit of which there are a number of parallel-connected light-emitting diodes. Each LED is in series with a zener diodes and a resistor. The zener diodes have dissimilar zener voltages as shown in the diagram. When the drop across R2 exceeds the sum of base-emitter voltage of T1, a zener voltage, and the threshold voltage of the LED in series with that zener diode, the relevant LED lights.

The diagram shows at which resistance value of the unit/component on test a particular LED lights. Bear in mind, however, that these values depend to some extent on the type of LED, and also that the zener voltages are subject to tolerances. Serious deviations may be corrected by the addition of a standard diode or a Schottky diode. It is also possible to add branches to individual requirements, or to use a bar display instead of LEDs. It is important that the op amp used has a rail-to-rail output since the input voltages as well as the output may rise to the peak supply voltage. This requirement is met by the MAX4322 as used in the prototype.

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Electric Guitar/Violin Preamplifier

Magnetic pick-ups in musical instruments have a relatively high output impedance. This can result in a reduction in treble response when connected via a long cable run or to equipment with a low input impedance. This preamplifier provides a high input impedance and a low impedance output, solving both issues. It has adjustable voltage gain and can run off a battery or DC plugpack. The input signal is AC-coupled to the non-inverting input of IC1a, part of a TL074 quad op amp. This has JFET input transistors and the input impedance is set by a 330kΩ bias resistor which also sets the DC level at this input to half supply (Vcc/2).

This is generated by a voltage divider comprising two 10kΩ resistors and bypassed by a 47µF capacitor to reject noise and hum. IC1a is configured as a non-inverting amplifier with a gain of between 2 and 20, depending on the setting of VR1. IC1a’s output is fed to VR2 via a 22µF capacitor, allowing the output volume to be set. The audio then passes to the non-inverting inputs of the remaining three op amps (IC1b-IC1d) which are connected in parallel to provide a low output impedance; it will drive a load impedance as low as 600Ω. The 100Ω resistors in series with the outputs provide short-circuit protection for the op amps and also prevent large currents from flowing between the outputs in case they have slightly different offset voltages.

The buffered signal is then AC-coupled to two output connectors using 47µF electrolytic capacitors. For Output 1, a 47kΩ resistor sets the output DC level to ground and a 220Ω series resistor provides further short-circuit protection. Output 2 is similar but includes another potentiometer (VR3) to allow its level to be set individually. Note that this means the impedance of Output 2 can be high (up to 2.5kΩ depending on the position of VR3’s wiper). The total harmonic distortion of this circuit is typically less than 0.01% with the gain set to six. If a TL064 is used instead of a TL074, the current drain will decrease but there will be more noise at the output. Finally, the input impedance can be increased by increasing the value of the 330kΩ resistor to suit high-impedance pick-ups.

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Phone Line Polarity Checker

Do you suspect your broadband speed (DSL) is slower than it should be? It could be as simple as incorrect polarity in your phone wall socket. Build this very cheap, very simple device to find out whether you need to change your wiring!

It might not seem that telephone line polarity is important, since the ring and voice signals sent over telephone lines are AC. However the lines are actually biased to 48V DC (less when in use) and so the polarity can matter. The main problem with incorrect polarity is that some DSL (Digital Subscriber Line) modems and routers can perform poorly in this circumstance. Since telephone wires are color-coded, it should be possible to simply check that the sockets are wired correctly. Unfortunately, there are multiple wiring colour schemes and they have changed over time. Believe it or not, the old color scheme (from around 15 years ago) is identical to the current colour scheme except that the polarity of both lines is reversed! This is why so many homes have this problem and yours may well be one of them.

The common telephone line is simply a copper pair, ie, two wires. As mentioned, there is usually a 48V DC bias across the pair which drops to around 8V when a telephone is “off-hook”. The ring voltage (around 90V AC) and the audio signal voltage (also AC) are overlaid on this DC bias. The DC power is “rectified” by each telephone on that line to run its own circuitry. Note, though, that this does not include cordless phones which usually use a plugpack, as their power requirements are far in excess of what the telephone line can deliver. (As an aside, that is the reason it is important to keep a line-powered telephone in your home so you can still make and receive calls if the mains power goes out.

Telephone exchanges can usually supply power from their backup batteries for up to some days, even if they are blacked out). Usually, telephone lines are run with 4-core cable. This allows up to two lines on the one cable. The first line is on the inner pair (pins 2 and 3) and the second line, if present, is on the outer pair (pins 1 and 4). Modern telephones use modular plugs, specifically RJ11 (6P2C, one line), RJ14 (6P4C, one or two lines) or RJ25 (6P6C, 1-3 lines). By the way, 6P4C stands for “six pins, four connectors”. Incidentally, “RJ12” connectors are physically compatible – and commonly available – so that is what we have used in this project.

Because modern phones rectify the DC voltage from the telephone lines before regulating it and because the ring and voice signals are AC, for voice communications the polarity doesn’t really matter.

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Gentle Battery Regulator

This small but very effective circuit protects a lead-acid battery (12-V solar battery or car battery) against overcharging by a solar module when the incident light is too bright or lasts too long. It does so by energizing a fan, starting at a low speed when the voltage is approximately 13.8 V and rising to full speed when the voltage exceeds 14.4 V (full-charge voltage). The threshold voltage (13.8 V) is the sum of the Zener diode voltage (12 V), the voltage across the IR diode (1.1 V), and the base-emitter voltage of the 2N3055 (0.7 V). In contrast to circuits using relays or IC amplifiers, the circuit has a gradual switching characteristic, which avoids relay chatter and the constant switching on and off near the switching point produced by a ‘hard’ switching point.

The circuit does not draw any current at all (auto power-off) below 13 V. Pay attention to the polarization of the Zener and IR diodes when building the circuit. The transistor must be fitted to a heat sink, since it becomes hot when the fan is not fully energized (at voltages just below 14 V). A galvanized bracket from a DIY shop forms an adequate heat sink. The indicated component values are for a 10-W solar module. If a higher-power module is used, a motor with higher rated power must also be used. The circuit takes advantage of the positive temperature coefficient of the lamp filament. The filament resistance is low at low voltages and increases as the voltage rises. This reduces the speed of the fan to avoid generating an annoying noise level. The lamp also provides a form of finger protection.

If you stick your finger into the fan blade, the lamp immediately takes over the majority of the power dissipation and lights brightly. This considerably reduces the torque of the fan. An ordinary 10-W or 20-W car headlight (or two 25-W headlights in parallel) can be used for the lamp. Don’t try to replace the LED by two 1N4001 diodes or the like, replace the ZPY12 by a ZPY13, or fit a series resistor for the LED. That would make the ‘on’ region too large.

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Sixties-Style 40W Audio Amplifier

In the early 1960s RCA brought out a transistor that was to become truly legendary: the 2N3055. With a pair of these devices, you could put together an audio power amplifier that could deliver a healthy 40 W into 8 Ω. The circuit described here is fully in tune with the spirit of that era. For example, there are only seven active components in each channel, which reflects the design simplicity typical of that era (and actually a timeless quality). This ‘retro’ power amplifier pumps 45 W into 8 Ω with an input signal level of 0.5 Vrms. It works as follows: the input signal is applied to the base of T1, while negative feedback from the output, attenuated by voltage divider R5/R6, is applied to the emitter of T1. The collector current of T1, which is proportional to the difference between the input and feedback signals, is fed to the base of T2.

Amplifier Circuit Diagram:

This transistor draws its operating current through R8 and R9 and provides voltage gain. Capacitor C6 is a bootstrap capacitor that hold the voltage across R9 nearly constant, so that the current through R9 is independent of the amplifier output signal level in the audio band. Transistors T4–T7 form a quasi-complementary push-pull output stage. In the early 1960s, there simply wasn’t any PNP transistor available that was truly complementary to the 2N3055. Designers came up with an ingenious way to get round this problem, which was to use a complementary Darlington pair consisting of a PNP driver transistor and an NPN power transistor. The schematic diagram clearly illustrates what is meant by a quasi-complementary push-pull output stage. Diode D1 provides balanced biasing for the output stage, which helps reduce distortion.

The operating point of the output stage is set and stabilized by transistor T3, which for this reason should be thermally coupled to the output transistors. The amplifier is powered from a single supply voltage at approximately 65 V, which is also ‘typical sixties’. Capacitor C1, with a value of 4700 μF, transfers the signal from the output stage to the load and provides a bit of protection for the speaker in case one of the transistors fails. The amplifier does not have output current limiting. Although this is not a critical shortcoming, a certain amount of caution is advisable. The only protection in this regard is provided by a slow-acting 1.6-A fuse in the supply line, which is intended to limit the damage if anything goes wrong.

Power Supply Circuit Diagram:

The power supply consists of a transformer, a bridge rectifier, four small capacitors and a 4700 μF electrolytic capacitor. This is enough to power a two-channel stereo amplifier. The LED is a power-on indicator and is intended to be fitted on the front panel. Assembling the circuit is very straightforward. Transistors T3, T4 and T5 should be fitted with heat sinks suitable for a TO126 package and with a thermal resistance less than 20 K/W. Transistors T2, T6 and T7 should all be fitted on a single heat sink with a thermal resistance of 2 K/W or less, using insulating washers and thermal paste. Before applying power to the circuit for the first time, set P2 to its maximum value, temporarily replace the fuse with a 47 Ω, 5 W resistor, and connect a voltmeter across R17.

Then switch on the power. The voltmeter should indicate 0 V. Now carefully adjust P2 until the voltmeter reads 15 mV, which corresponds to a quiescent current of 50mA. Then switch off the power and install the fuse in place of the power resistor. After this, check the voltage across R17 again (with the power on) and if necessary adjust it to 15mV. This is fun DIY project, cheap and unpretentious. Nevertheless, the sound quality of this amplifier is respectable. The distortion level gives no grounds for complaint. Of course, it’s not a figure with an incredible number of zeros after the decimal point, but the idea here is to brush up on sixties technology.



Author: Joseph Kreutz (Germany) - Copyright: Elektor Electronics 2011

 

 

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Voltage Limiter For Guitar Amplifiers

Guitar amplifiers using output devices such as the TDA7293 (100 W) or LM3886 (68 W) are surprisingly often damaged as a result of excessive supply voltage in the quiescent state. The transformers are often used so close to their specification that this problem can even be caused by a high mains input voltage. In most countries the domestic AC outlet voltage is permitted to rise as high 10 % above the nominal (published) value. Since replacing the transformer is not an attractive proposition, the author developed a relatively simple electronic solution to the overvoltage problem: a voltage limiter for the symmetric supply to the amplifier. The circuit is based on the classical voltage regulator arrangement of a Zener diode connected to the base of a pass transistor.

However, in this version we replace the conventional bipolar transistor with a power MOSFET. The circuit is symmetrical with respect to the negative and positive supplies, and so we shall only describe the positive half. The input voltage (at most 50 V) supplies the chain of Zener diodes D1, D2 and D3 via resistor R3. The resistor limits the current through the Zener diodes to about 5mA. The series connection of Zener diodes has the advantage that their dissipation is divided among them, as well as giving more options for the total voltage drop by judicious selection of individual components. The sum of the diode voltages (39 V with the values given) must be greater than the desired limiting output voltage by the gate-source voltage of the MOSFET.

Circuit diagram:

C1 smooths the voltage across the Zener diode chain. The circuit therefore not only limits the voltage, but also reduces the ripple (hum component) on the supply. The gate of the HEXFET is driven via R1. In conjunction with C4, this prevents the FET from oscillating. Without any load the output voltage is rather higher than expected. With a small load, such as that presented by the output stage in its quiescent state, it falls to the desired value. The circuit then does not provide regulation of the output voltage, but rather a stabilization function. The operation of the negative half of the circuit is identical to that of the positive half apart from the polarity of the voltages, and hence a P-channel MOSFET must be used there.

It is worth noting that there is a relatively large degree of variation (up to a few volts) in the gate-source voltage of the HEXFETs used. This can be compensated for by selecting the Zener diodes in the chain and the current through them, but for most applications the exact voltage at which limiting begins to occur will not be critical. The HEXFETs must be provided with adequate cooling. If possible, they can be attached to the heatsink already present in the amplifier; otherwise, a separate heatsink will be required. A thermal rating of 2.5 K/W will be suitable.



Author: Alfred Rosenkränzer (Germany) - Copyright: Elektor Electronics 2011

 

 

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